FIG. 1 shows a block diagram of a generic wireless transceiver, in which a processing circuit 10, such as, for example, a digital signal processor (DSP), supplies a baseband (BB) transmitting signal TXBB. The above baseband transmitting signal TXBB is converted by a transmitter circuit 20 into a RF transmitting signal TXRF. For instance, typically the transmitter circuit 20 comprises a modulator, such as, for example, a mixer or an analog multiplier, which modulates the signal TXBB with a high-frequency carrier signal LO. In addition, the transmitter circuit may also comprise filters, amplifiers, etc. Finally, the transmitting signal TXRF is sent to at least one antenna 30.
In a complementary way, an RF receiving signal RXRF received via the antenna 30 is converted via a receiver circuit 40 into a baseband receiving signal RXBB. For instance, typically the receiver circuit 40 comprises a demodulator, such as, for example, a mixer, which demodulates the signal RXRF using the carrier frequency LO. Also, the receiver circuit may comprise filters, amplifiers, etc. For instance, the carrier signal LO may be supplied by an oscillator or synthesizer 50.
A particular architecture of the receiver 40 is the architecture of a so-called “low-IF” type. Basically, in a low-IF receiver, the RF signal RXRF received is demodulated at a lower, non-zero, frequency, the so-called “intermediate frequency” (IF), which typically ranges from hundreds of kilohertz (kHz) to some megahertz (MHz).
Receivers with a low-IF architecture are commonly used in transceiver systems on account of their relatively low complexity and robustness. The main characteristic of this architecture is the fact that the RF signal RXRF received is converted by means of a system of a heterodyne type to a significantly lower frequency, hereinafter designated by fIF. In particular, the heterodyne system is implemented through a mixer that carries out multiplication of the RF signal by an ideally pure tone (LO) with frequency fLO, appropriately generated by the synthesizer 50 in such a way that:fIF=fRF−fLO.  (1)The high-frequency components generated by the multiplication can be subsequently filtered along the receiving chain.
The choice of the frequency fIF has a considerable effect on the design of the analog system in so far as, if it is sufficiently high, it enables reduction of the problems of flicker noise and DC offsets generated by the chain of receiver circuits. On the other hand, an excessive increase of the frequency fIF may lead to an increase of the power dissipation of the analog-to-digital converter (ADC) and also of the DSP in so far as it requires a higher working frequency.
Low-IF receivers normally use in-phase quadrature signals (i.e., of a complex-envelope type) both to facilitate demodulation thereof and to solve the problem of image rejection. The in-phase quadrature signals are periodic waveforms that have a phase difference equal to one quarter of their period, namely, 90°. Consequently, as highlighted in FIG. 2, the low-IF receiver circuit 40 receives at input the RF receiving signal RXRF. In this example, the signal RXRF is amplified via an amplifier 402, such as, for example, a low-noise amplifier (LNA).
In particular, in the case where the receiver 40 operates with signals I and Q that are in quadrature with respect to one another, the amplified signal, i.e., the signal at output from the amplifier 402, is sent to two branches: a first branch for the in-phase (I) component, and a second branch for the quadrature (Q) component. In this case, each branch comprises a demodulator 404, such as, for example, a mixer, which carries out multiplication of the RF signal by respective signals LOI and LOQ, and a filter 406, which, by filtering the high-frequency components, yields the evolution in time of the respective component IRX and QRX.
To interface those signals with the processing circuit 10, respective analog-to-digital (A/D) converters 408 may be provided. Consequently, reception of a complex signal calls for generation, upstream, of the in-phase signal LOI and the quadrature signal LOQ, i.e., having a phase shift of 90° with respect to one another. Generation of the tones LOI and LOQ with controlled phase shift calls for an accurate design of the circuit 50 that will limit as far as possible the inevitable cumulative phase errors.
The techniques normally employed envision use of an oscillator inserted in a phase-locked loop (PLL), where the oscillator may, for example, be a voltage-controlled oscillator (VCO) 502, and multiphase filters or frequency dividers 504; the latter approach, however, envisions generation of a tone by the synthesizer, the frequency of which should be at least twice the desired one. Against this disadvantage, the active division circuit enables introduction of techniques for control of the phase error that can also compensate for possible phase errors accumulated in the receiving chain.
The precision on the amplitude and phase of the in-phase quadrature signals I and Q may be important in RF communication systems that adopt in reception of the low-IF architecture, since it affects the levels of performance of the receiver. The low-IF architecture, which yields benefits from the standpoint of offset and flicker noise, presents in fact an image signal that may be very close to the channel of interest and that hence may require use of two in-phase quadrature signals for implementing rejection of the image signal.
A typical problem of the low-IF receiver may include so-called “image response or rejection.” With reference to FIG. 3a, the problem may include the fact that a generic heterodyne system produces a frequency conversion both of the desired channel CHN. In this case, this is at a frequency fCHN=fLO+fIF, and of its image IMG positioned at fIMG=fLO−fIF, which at this point can hardly be rejected with a classic real analog filter, such as, for example, the filter 406, in so far as both of the channels are brought to the frequency fIF, since the components CHN and IMG come to be superimposed during demodulation in the demodulators 404 (see FIG. 3b).
Selection of the channel CHN may in any case be made by complex-filtering techniques, which can be implemented either in an analog or in a digital way and operate on the complex (in-phase quadrature) signal received by selecting the desired channel CHN from the image IMG and from other possible out-of-band interfering signals. The effectiveness of the complex filter in rejection of the image IMG is, however, markedly affected by the phase and amplitude mismatch or errors that accumulate on the in-phase quadrature signals at input, where the phase mismatch is defined as the deviation with respect to the 90° phase shift expected between the signals I and Q, and the amplitude mismatch is defined as the lack of amplitude correspondence between the signals I and Q. The image-rejection ratio is described, for example, in the paper by Q. Gu, “RF System Design of Transceivers for Wireless Communications,” New Work, USA, Springer, 2005 (Gu reference).
For example, FIG. 4, which is disclosed in the Gu reference, shows a typical relation for image rejection (IR) with respect to the phase mismatch, or “Phase Imbalance”, as appears on the horizontal axis, and the amplitude mismatch, or “Amplitude Imbalance,” as appears on the vertical axis. The errors of the in-phase quadrature signals I and Q are correlated to the image-rejection (IR) ratio. The relation that expresses the image rejection IR with respect to the phase mismatch φ and the amplitude mismatch δ may be expressed via the following equation:
                              IR          =                      10            ⁢                                                  ⁢            log            ⁢                                          1                +                                  2                  ⁢                                      (                                          1                      +                      δ                                        )                                    ⁢                  cos                  ⁢                                                                          ⁢                  φ                                +                                                      (                                          1                      +                      δ                                        )                                    2                                                            1                -                                  2                  ⁢                                      (                                          1                      +                      δ                                        )                                    ⁢                  cos                  ⁢                                                                          ⁢                  φ                                +                                                      (                                          1                      +                      δ                                        )                                    2                                                                    ;                            (        2        )            where δ is the amplitude error (expressed in decibels) and φ is the phase error with respect to the ideal 90° phase shift between the two signals I and Q. Consequently, normal techniques of correction of the errors are introduced in such a way as to maximize the image rejection that can be obtained in accordance with the specifications of the system and with the effective selectivity of the complex filter.
For instance, the technique described in Li Yu, W. Martin Snelgrove, “A Novel Adaptive Mismatch Cancellation System for Quadrature IF Radio Receivers”, IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: ANALOG AND DIGITAL SIGNAL PROCESSING, VOL. 46, NO. 6, JUNE 1999, is a known technique because it operates digitally on the complex signal received, producing a simultaneous correction of amplitude and phase mismatch prior to complex filtering.
Alternatively, the technique described in Oscar Stella. “Automatic In-phase Quadrature Balancing AIQB,” October 2006 (Rev C: Jul. 10, 2012) may be used, where a mismatch correction is made by correlating appropriately different harmonic contributions of the signal received. This approach calls, however, for an operation of a Fast Fourier Transform (FFT), which is typically more burdensome from the computational standpoint.
However, the increasing demand for low-consumption systems clashes with producing high-performance ADC circuits, which in general prove particularly burdensome from the consumption standpoint and frequently force the digital circuitry to operate at higher sampling frequencies, thus weighing even more heavily on the power budget. Optimization of the circuits and appropriate distribution of the functions linked to selection of the channel CHN within the low-IF architecture may, however, contribute significantly to the reduction of the overall consumption of the system, reducing in particular the performance required of the ADC and the digital circuitry.
In this sense, the architecture proposed in FIG. 5 includes a complex filter 412 of an analog type upstream of the A/D conversion. The complex filter 412 is ideally able to select the desired channel CHN from any other interfering channel (including the image IMG), intrinsically limiting the band requirement and the resolution of the ADC and hence also the consumption of the processing unit 10.
Elimination of the image channel IMG, moreover, enables for some specific modulation formats demodulation of the channel received without necessarily having a complex signal, and consequently it is possible to eliminate also one of the two A/D converters 408. Against the advantages set forth there may remain the problem of correction of the phase and amplitude errors at input to the complex filter 412, since in the presence of the filter and of an A/D converter of limited performance it may not be possible to use the techniques proposed by Li Yu and Oscar Steila.
In the example of FIG. 5, the correction is made upstream of the complex filter 412, by appropriately acting on the frequency dividers present within the generator 50, for example, in the circuit 504, and on the amplitude error at the baseband level, for example, by adding a respective amplifier with configurable amplification coefficient 410, for example, between the filter 406 and the filter 412. There may also exist different approaches that envision making the correction of both of the errors by acting only on the I/Q components received at IF frequency.